System for processing digitized communication signals provided over plural RF channels

ABSTRACT

A wireless digital subscriber communications system includes a base station and a subscriber unit and uses communications signals provided over a plurality of radio frequency channels. A processor provides an output phase signal corresponding to a selected output digital frequency, and a lookup table is used having two sets of predefined stored values pertaining to the amplitude of a signal for a single quadrant. In particular embodiments, predefined stored values include coarse angle approximations and fine angle approximations and a sine and cosine generator receives the phase signal and generates sine and cosine waveforms utilizing amplitude values obtained from the lookup table. In a further embodiment, the phase signal includes phase data and specifies the quadrant and the algebraic sign of the phase data, with the sine and cosine generator accessing the lookup table differently depending upon the quadrant and sine of the phase data, such that the lookup table provides an amplitude value from the sets of predefined stored values based on the phase data. A modulator combines the sine and cosine waveforms to produce the selected output digital frequency and modulates digital frequency.

BACKGROUND

[0001] This application is a continuation of U.S. application Ser. No.10/412,456, filed Apr. 11, 2003, which is a continuation of U.S.application Ser. No. 10/223,750, filed Aug. 20, 2002, now U.S. Pat. No.6,587,516, issued Jul. 1, 2003, which is a continuation of U.S.application Ser. No. 09/593,307, filed Jun. 13, 2000, now U.S. Pat. No.6,449,317, issued on Sep. 10, 2002, which is a continuation of U.S.application Ser. No. 09/228,140, filed Jan. 11, 1999, now U.S. Pat. No.6,078,629, issued on Jun. 20, 2000; which is a continuation of U.S.application Ser. No. 08/881,339, Jun. 24, 1997, now U.S. Pat. No.5,859,883, issued on Jan. 12, 1999; which is a continuation of U.S.application Ser. No. 08/587,008, filed Jan. 11, 1996, now U.S. Pat. No.5,694,430, issued Dec. 2, 1997; which is a continuation of U.S.application Ser. No. 08/445,082, filed May 22, 1995, now U.S. Pat. No.5,644,602, issued Jul. 1, 1997; which is a divisional of applicationSer. No. 08/222,670, filed Apr. 4, 1994, abandoned; which is acontinuation of U.S. application Ser. No. 07/940,662, filed Sep. 4,1992, now U.S. Pat. No. 5,325,396, issued Jun. 28, 1994; which is acontinuation of U.S. application Ser. No. 07/658,065, filed Feb. 20,1991, Now U.S. Pat. No. 5,146,473, issued Sep. 8, 1992; which is acontinuation of U.S. application Ser. No. 07/394,497, filed Aug. 14,1989, now U.S. Pat. No. 5,008,900, issued Apr. 16, 1991, which areincorporated by reference as if fully set forth.

FIELD OF INVENTION

[0002] The present invention generally pertains to subscribercommunications systems and is particularly directed to an improvedsubscriber unit for wireless communication with a base station in awireless digital subscriber communication system.

[0003] A typical subscriber unit is described in U.S. patent applicationSer. No. 06/893,916 filed Aug. 7, 1986 by David N. Critchlow et al. NowU.S. Pat. No. 4,825,448. A base station used with such a subscriber unitin a wireless digital subscriber communication system is described inU.S. Pat. No. 4,777,633 to Thomas E. Fletcher, Wendeline R. Avis,Gregory T. Saffee and Karle J. Johnson. The subscriber unit described inU.S. Pat. No. 4,825,448 includes means for transcoding a digital voiceinput signal to provide digital input symbols; means for FIR filteringthe digital input symbols; means for deriving an analog intermediatefrequency input signal from the filtered input symbols; means forcombining the intermediate frequency input signal with an RF carrier forradio transmission to the base station; means for demodulating an outputsignal received from the base station to provide digital output symbols;and means for synthesizing a digital voice output signal from thedigital output symbols. The subscriber unit includes a basebandprocessor chip and a modem processor chip. Both are TMS32020 digitalsignal processors. The baseband processor chips perform the transcodingof the digital voice input signal, the synthesis of the digital outputsymbols, and various baseband control functions; and the modem processorchip performs the FIR filtering of the digital input symbols, and thedemodulation of the output signal received from the base station. Themodem processor chip generally acts as the master for the system.

SUMMARY OF THE INVENTION

[0004] A wireless digital subscriber communications system includes abase station and a subscriber unit and uses communications signalsprovided over a plurality of radio frequency channels. A processorprovides an output phase signal corresponding to a selected outputdigital frequency, and a lookup table is used having two sets ofpredefined stored values pertaining to the amplitude of a signal for asingle quadrant. In particular embodiments, predefined stored valuesinclude coarse angle approximations and fine angle approximations and asine and cosine generator receives the phase signal and generates sineand cosine waveforms utilizing amplitude values obtained from the lookuptable. In a further embodiment, the phase signal includes phase data andspecifies the quadrant and the algebraic sign of the phase data, withthe sine and cosine generator accessing the lookup table differentlydepending upon the quadrant and sine of the phase data, such that thelookup table provides an amplitude value from the sets of predefinedstored values based on the phase data. A modulator combines the sine andcosine waveforms to produce the selected output digital frequency andmodulates digital frequency.

BRIEF DESCRIPTION OF THE DRAWINGS

[0005]FIG. 1 is a block diagram of a preferred embodiment of thesubscriber unit of the present invention.

[0006]FIG. 2 is a block diagram of the FIR chip included in theembodiment shown in FIG. 1.

[0007]FIG. 3 is a block diagram of the DIF chip included in theembodiment shown in FIG. 1.

[0008]FIG. 4 illustrates the processing tasks performed by the processorchip shown in the embodiment of FIG. 1.

[0009]FIG. 5 illustrates the processing routines included in modemprocessing task shown in FIG. 4.

DEFINITION OF ABBREVIATIONS AND ACRONYMS

[0010] The following is a definition of abbreviations and acronyms usedherein: A/D Analog to Digital AGC Automatic Gain Control ASICApplication Specific Integrated Circuit BPSK Binary Phase Shift KeyingCCT Channel Control Task CCU Channel Control Unit CRC Cyclic RedundancyCheck DAC Digital to Analog Converter DDS Direct Digital Synthesizer DIFDigital Intermediate Frequency DIP Dual In-line Package DOR Data OutputReady DPSK Differential Phase Shift Keying DSP Digital Signal ProcessingEPROM Erasable Read Only Memory FIR Finite Impulse Response I/OInput/Output LSB Last Significant Bit MPT Modem Processing Task MSB MostSignificant Bit MUX Multiplexer PCM Pulse Code Modulation PLL PhaseLocked Loop PWM Pulse Width Modulation QPSK Quadrature Phase ShiftKeying RAM Random Access Memory RCC Radio Control Channel RELP ResidualExcited Linear Predictive RF Radio Frequency ROM Read Only Memory RXReceive RXCLK Receive Clock RXSOS Receive Start of Slot SCT SubscriberControl Task SLIC Subscriber Line Interface Circuit SPC SignalProcessing Control SPT Signal Processing Task SPTCTL Signal ProcessingTask Controller SSB Switch-hook Sample Buffer TDM Time DivisionMultiplexing TX Transmit TXCLK Transmit Clock UART UniversalAsynchronous Receiver Transmitter VLSI Very Large Scale Integration XORExclusive Or

DESCRIPTION OF THE PREFERRED EMBODIMENT

[0011] Referring to FIG. 1, a preferred embodiment of the subscriberunit of the present invention includes a telephone interface circuit 10,a SLIC and codec circuit 11, a processor chip 12, a fast memory 13, aslow memory 14, an address decoder 15, a FIR chip 16, a DIF chip 17, aDAC 18, an A/D converter 19, a radio 20, a ringer circuit 21, and anoscillator 22.

[0012] The FIR chip 16, which is an ASIC chip, is interfaced with theDIF chip 17 by lines 23 and 24, to the processor chip 12 by processorbus 25 and line 26, to the A/D converter 19 by line 27, to the SLIC andcodec circuit 11 by line 29, to the radio 20 by line 30, and to theringer circuit 21 by line 31.

[0013] The telephone interface circuit 10 is interfaced with a telephone32, which converts sound waves into an input voice signal, and convertsan output voice signal into sound waves.

[0014] The SLIC and codec circuit 11 is coupled to the telephoneinterface circuit 10 for converting the input voice signal into abaseband digital input signal, which is provided to the processor chip12.

[0015] In an alternative embodiment (not shown), the processor chip isalso interfaced directly with an UART for alternatively receivingdigital input signals directly from and sending digital output signalsdirectly to a digital signal I/O device.

[0016] The processor chip 12 includes a model TMS320C25 digital signalprocessor, which transcodes the baseband digital input signal inaccordance with a RELP algorithm to provide TX data digital inputsymbols on the processor bus 25. The use of a digital signal processorto perform a RELP algorithm is described in International PatentApplication No. PCT/US85/02168, International Publication No. WO86/02726, published May 9, 1986.

[0017] The FIR chip 16 FIR filters the digital input symbols andprovides I,Q data to the DIF chip 17 on lines 24.

[0018] The DIF chip 17 interpolates the filtered digital input symbols,and modulates a digital intermediate frequency signal with theinterpolated input symbols to provide a modulated digital input signal.

[0019] The DAC 18 converts the modulated digital input signal into amodulated analog input signal.

[0020] The radio 20 transmits the modulated analog input signal to thebase station; and receives and demodulates a modulated analog outputsignal from the base station.

[0021] The oscillator 22 is a free running oscillator, that providesclock signals for the processor chip 12.

[0022] A description of the relationship between the subscriber unit andthe base station is contained in U.S. Pat. No. 4,777,633.

[0023] The A/D converter 19 converts the demodulated received analogoutput signal into a digital output signal containing digital outputsymbols.

[0024] The processor chip 12 synthesizes a baseband digital outputsignal from the digital output symbols. Synthesis of RELP transcodedsymbols by a digital signal processor also is described in InternationalPublication No. WO 86/02726. The processor chip 12 further performs echocancellation as described in U.S. Pat. No. 4,697,261 to David T. K. Wangand Philip J. Wilson.

[0025] The SLIC and codec circuit 11 converts the baseband digitaloutput signal into the output voice signal that is provided by thetelephone interface circuit to the telephone 32.

[0026] The FIR chip 16 consolidates circuit functionality into a VLSIdevice in order to reduce production cost of the subscriber unit byeliminating many separate medium scale integration parts.

[0027] Referring to FIG. 2, the FIR chip 16 includes a fanout buffer 33,an internal decoding module 34, an RX sample buffer 35, control andstatus registers 36, an external address decoding module 37, a watchdogtimer module 38, an RX timing module 39, a TX timing module 40, a TX FIRfilter 42, a codec timing module 44, and a ringer control module 45.

[0028] The FIR chip 16 provides 45 millisecond frame marker generation,11.25 millisecond slot marker generation, 16 KHz symbol clockgeneration, timing adjustment circuits, RX sample buffering, TX symbolbuffering, 8 KHz codec timing generation, processor interface decoding,ringer timing generation, external address decoding and watchdog timerreset generation. The FIR chip 16 buffers two 5-bit TX symbols at a 8KHz rate. The FIR chip 16 converts and filters the TX symbols into I andQ data symbols, with each such symbol being 10-bits at a rate of 160KHz. The I and Q data are interleaved and output to the DIF chip 17 at arate of 320 KHz. The FIR chip 16 also buffers RX data samples at a 64KHz rate; and four RX data samples are read by the processor chip 12 ata 16 KHz rate. Timing clocks and signals are generated by the FIR chip16 from an incoming 3.2 MHz master clock signal. The processor chip 12is synchronized to these data rates by slot and symbol interruptsgenerated by the FIR chip 16. The codec and processor 8 KHz timingstrobe and codec clock are generated by the FIR chip 16 and synchronizedto the time of the incoming RX samples. The FIR chip 16 also generatescontrol and timing signals for controlling the shape and timing of theringing voltage provided by the ringer circuit 21. The watchdog timermodule 38 provides a reset signal in the event that the processor chip12 does not execute instructions properly.

[0029] The fanout buffer 33 buffers a 3.2 MHz master clock signalreceived on line 23 a from the DIF chip 17, an advanced 3.2 MHz clocksignal received on line 23 b from the DIF chip 17, and a reset signalreceived on line 51 from the watchdog timer 38. Unless otherwiseindicated, all timing within the FIR chip 16 is derived from the 3.2 MHzclock signal on line 23 a. The advanced 3.2 MHz clock signal on line 23b leads the 3.2 MHz clock signal on line 23 a by one cycle of a 21.76MHz reference signal that is present within the DIF chip 17. The 3.2 MHzclock signal is derived from the 21.76 MHz reference in the DIF chip 17and the minimum pulse width is therefore 276 nanoseconds. The advanced3.2 MHz clock signal from line 23 b is provided from the buffer 33 viainternal line 47 to the TX FIR filter 42, and the codec timing module44. The TX FIR filter 42 is implemented in part by a ROM, which ispseudo-static and requires its enable input to be deactivated by theadvanced 3.2 MHz clock signal on line 47 between successive accesses.

[0030] The HW reset signal on line 51 resets all internal circuits ofthe FIR chip 16 and provides a hardware reset to the modules of FIG. 1.

[0031] The internal clocks are either buffered versions of the 3.2 MHzmaster clock signal received on line 23 a or divisions of this clock.

[0032] The internal address decoding module 34 allows the processor chip12 to access the internal functions of the FIR chip 16 for the purposeof controlling such functions and determining their status. The internaladdress decoding module 34 receives processor addresses and processorstrobes on bus 25. The internal address decoding module 34 providesoutput signals on internal bus 48.

[0033] The output signals on bus 48 from the internal address decodingmodule 34 include a read enable signal to the RX sample buffer 35, acontrol write signal and status read signals to the control and statusregisters 36, a write signal to the TX FIR filter 42, slot and clockwrite signals to the RX timing module 39, a write signal to the TXtiming module 40, and control signals to the TX FIR filter module 42 andthe RX sample buffer 35, and an AM Strobe signal, which causes the RXtiming module 39 to reset slot timing. Only one of the respective reador write signals on bus 48 from the internal address decoding module 34is active at any one time.

[0034] The RX sample buffer 35 receives four samples for each RX symboltime from the A/D converter 19 via line 27 a at a 64 KHz rate; buffersup to two symbols of data, which is eight samples total; and then sendssuch data samples to the processor chip 12 via the processor bus 25. TheRX sample buffer 35 is implemented in a dual-page RAM. The RX samplebuffer 35 receives a read enable signal on internal bus 48 from theinternal address decoding module 34 and a write strobe signal oninternal line 49 from the RX timing module 39.

[0035] The control and status registers 36 allow the processor chip 12to control the internal functions of the FIR chip 16, and allow theprocessor chip 12 to read the status of the TX FIR filter 42 and RXsample buffer 35, and other internal signals. The control signals areprovided by the processor chip 12 via the processor bus 25 and thestatus indications are derived from various internal modules of the FIRchip 16. The status indications are provided to the processor chip 12via the processor bus 25. The status indications are RX Underrun RXOverrun, TX Underrun, TX Overrun, Start-of-Frame, RX Start of slot, TXsymbol Clock, RX Symbol Clock and TX FIR filter Overflow. The controlsignals, which are provided by the control registers 36 to the internalcircuits via the internal bus 48, include the following: TX Enable,Modulation Level, Ringer Enable, Software Reset, Tristate, and WatchdogStrobe.

[0036] The TX Enable signal indicates the beginning of a TX slot basedupon the TX delay established in the TX timing module 40.

[0037] The Modulation Level signal is provided to the RX timing module39 and determines whether a slot length is 180 or 360 symbols.

[0038] The Software Reset signal allows the processor chip 12 to resetinternal functions within the FIR chip 16.

[0039] The Tristate signal allows the processor chip 12 to disable theoutputs of the FIR chip 16.

[0040] The Ringer Enable signal allows the processor chip 12 to turn theringer circuit 21 on and off. This signal provides a two-second andfour-second cadence for the ringing signal.

[0041] The Watchdog Strobe allows the processor chip 12 to reset thewatchdog timer module in order to keep a hardware reset from occurring.

[0042] The processor chip 12 receives a RX clock interrupt (RXCLKINT)signal from the RX timing module 39 via line 26 c when data has beenwritten into the first four locations of the dual-page RAM of the RXsample buffer 35. The processor chip 12 then reads the RX samples fromthe first four locations of the dual-page RAM via processor bus 25. Atthis time samples are being written into the next four locations of thedual-page RAM at a 64 KHz rate. The 16 KHz event is a derivative of the64 KHz event, which keeps the read and write events synchronized. Thisensures that read and write operations do not occur at the same time atany one memory location and also ensures adequate response time from theprocessor chip 12.

[0043] A TX symbol buffer in the TX FIR filter 42 receives TX symbolsfrom the processor chip 12 via the processor bus 25 and buffers up totwo TX symbols. The processor chip 12 is interrupted every other TXsymbol time to write two more symbols into the TX symbol buffer.

[0044] The TX symbol buffer in the TX FIR filter 42 receives a writesignal via the internal bus 48 from the internal address decoding module34.

[0045] After each TX clock interrupt (TXCLKINT) signal at 8 KHz on line26 a, the processor chip 12 writes out two 5-bit TX symbols. The data isin a DPSK gray code format. The TX symbol buffer outputs a symbol every16 KHz for processing by the TX FIR filter 42. This data is doublebuffered due to an asynchronism between the FIR chip 16 and theprocessor chip 12. The last data value is repeated until new data iswritten. Null data can be repeated in this manner. The TX symbol bufferis cleared during a reset.

[0046] During training, a fixed sequence of symbols is sent to the FIRchip 16 by the processor chip 12. The FIR chip 16 performs FIR filteringon these symbols and outputs I,Q pairs to the DIF chip 17.

[0047] The radio 20 loops the data back to the AID converter 19. Thesamples are read by the processor chip 12 as in the on-line mode and thecoefficients of the processor RX filter implemented in the processorchip 12 are adjusted. The only timing critical for training is generatedby the RX and TX timing modules 39, 40.

[0048] The RX timing module 39 generates all reference clocks andstrobes for processing the RX symbols. The timing is adjusted by theprocessor chip 12 so that processing can be synchronized to the RXsamples received via line 27 a from the base station. The RX timingmodule 39 includes an RX clock fractional timing circuit and an RX Slottiming circuit. The purpose of these two circuits is to synchronize themodem receive timing within the processor chip 12 to the RX samplesreceived on line 27 a from the base station, and via the A/D converter19, and also to regulate the TX timing module 40 and the codec timingmodule 44.

[0049] The RX timing module 39 is clocked at a 3.2 MHz rate and receivesthe following control signal inputs from the processor chip 12 via theprocessor bus 25: an AM Strobe signal, an RX Slot Clock Write signal,and an RX Bit Tracking signal.

[0050] Several outputs are generated by the RX timing module 39. A 64KHz write strobe is provided on line 49 to control writing to the RXsample buffer 35. A 64 KHz A/DSYNC strobe signal is provided on line 27b to the A/D converter 19 to synchronize the operation thereof. A 8 KHZstrobe signal also is provided to the codec timing module 44 via line52. A 16 KHZ RX clock interrupt (RXCLKINT) signal on line 26 c and RXstart-of-slot interrupt (RXSOSINT) signal on line 26 b are output to theprocessor chip 12. A pre-RX slot timing strobe is provided on line 54 tocontrol the TX timing module 40.

[0051] The fractional timing circuit in the RX timing module 39 is setby the processor chip 12 to generate the RX start of slot interruptsignal on line 26 b. The processor chip 12 determines the location of anAM hole (strobe signal) transmitted by the base station duringacquisition. When the processor chip 12 detects the AM strobe signal,the slot timing circuit in the RX timing module 39 is reset by a resetsignal from the processor chip 12. This aligns the frame and slotmarkers to the AM strobe signal. The frame marker is a 62.5 μsec pulseoccurring every 45 milliseconds. The slot marker is a 62.5 μsec pulserepeating every 11.25 millisecond, or 22.5 milliseconds when in a QPSKmode.

[0052] The incoming RX symbols are demodulated by the processor chip 12and timing is further adjusted if necessary. To adjust the 16 KHz RXsymbol clock the processor chip forces the fractional timing (bittracking) circuit to shorten or lengthen the 64 KHz strobe by up tofifty 3.2 MHz cycles.

[0053] The processor chip 12 monitors the relationship of the RX symbolsto the frame timing and makes adjustments to the 16 KHz RX clockaccordingly. When the RX clock is adjusted the slot and frame markersare changed also because they are a derivative of the RX clock.

[0054] To keep the number of Pulse Code Modulated (PCM) samples providedto and from the SLIC and codec circuit 11 synchronized to the frametiming, the RX timing module 39 controls the codec timing module 44.

[0055] The TX timing module 40 includes a TX delay circuit and a TXcontrol timing circuit. These circuits generate a TX clock interrupt(TXCLKINT) signal which is provided to the processor chip 12 via line 26a. The TX timing module 40 is synchronized to the RX timing module 39 bythe pre-RX slot timing strobe, which is provided to the TX timing moduleby the RX timing module 39 on line 54 and used to reset the TX delaycircuit, which in turn generates the TX slot marker. Timing of the TXclock is based on the internal 3.2 MHz clock.

[0056] The processor chip 12 also controls the TX delay and TX timingcircuits by providing TX data write control signals over the processorbus 25. The TX timing module 40 provides a T/R control signal on line 30to the radio 20. This signal determines whether the radio istransmitting or receiving data.

[0057] The TX timing module 40 also controls TX symbol shifting, ROMaddressing, accumulation timing, and I,Q product storage for output tothe DIF chip 17.

[0058] The TX timing module 40 provides control signals on line 56 forkeeping the TX FIR filter 42 synchronized to the TX symbol and slottiming. Such synchronization is accomplished in accordance with the TXslot timing marker. After a reset, the TX timing module 40 activelygenerates control signals onto line 56 once a TX slot begins.

[0059] The TX FIR filter 42 module includes a ROM, which implements aFIR filter by providing I and Q data products in response to the ROMbeing addressed for lookup by a combination of TX symbols received fromthe processor chip 12 via the processor bus 25 and SINE and COSINEcoefficient counts provided by a counter within the TX FIR filter module42. The TX FIR filter 42 accumulates six sequential I and Q dataproduces and stores results for output to the DIF chip 17 via line 24 a.

[0060] The minimum frequency required for operation of the TX FIR filter42 is determined by the symbol rate (16 KHz) times the number of I and Qsamples (2) times the number of coefficients (10) times the number oftaps (6)=1.92 MHz. The master clock of 3.2 MHz meets this minimumfrequency requirement. Wait periods are added to compensate for thefaster execution time.

[0061] The TX timing module 40 is clocked at a 3.2 MHz clock rate, whichdefines one state period. Because this clock rate is greater than therequired minimum of 1.92 MHz the TX FIR filter 42 generates signals forthe first six out of ten state periods.

[0062] Each new TX symbol must be loaded into a circular buffer in theTX FIR filter 42 at the rate of 16 KHz. The new TX symbol and theprevious five TX symbols are stored in the circular buffer. The oldestTX symbol is dropped when a new TX symbol is shifted in. The TX FIRfilter 42 output rate is 320 KHz. From each TX symbol, ten I data valuesare generated and ten Q data values are generated. Table 1 below showshow I, Q and null information can be derived from each 5-bit value.TABLE 1 BIT 1 BIT 2 BIT 3 BIT 4 BIT 5 I & Q LSB I & Q I MSB Q MSB NULL

[0063] The data in the circular buffer is rotated every 6 out of 10states. One new TX symbol and the five previous TX symbols reside in thecircular buffer for twenty of these ten state periods. The coefficientportion of the ROM address is also increased every six out of ten stateperiods. An accumulator in the TX FIR filter 42 adds the results of eachI-data product provided from the ROM for each of the six state periods.Therefore the accumulator register is cleared for the first addition,and each successive addition result is clocked into a feed back registerof the accumulator so it can be added to the newly looked-up product.Once six additions occur the result is clocked into an output shiftregister. The same process occurs for the same coefficients and theQ-data products provided from the ROM for each TX symbol.

[0064] The ROM address lines allow sixty COS coefficient and sixty SINcoefficient lookups for four possible I,Q data indexes. This requiresseven address lines for coefficients and two address lines for I,Q data.The output of the FIR filter requires 10 bits. Two extra bits arerequired to maintain accuracy of the fractional portion of the lookupvalue. This makes the ROM size 512.times.12. The MSB of the I,Q dataindex is passed around the ROM to a 1's complement circuit which forcesthe output of the ROM to be inverted or not inverted.

[0065] If the symbol addressing the ROM is a null symbol the null bitcontrols four of the seven coefficient address lines. Since sevenaddress lines are used for coefficient lookup this provides 128locations. Only 120 coefficients are needed. This leaves eight unusedlocations. Zero values are stored in these locations so null informationcan be easily output from the ROM.

[0066] A 2's complement function is implemented by using a 1'scomplement and carrying in a logic 1 in the succeeding adder. The outputof the adder is wrapped around to the input of the adder for successiveadditions or output through a MUX to an output shift register. Theoutput is rounded off by using only the ten upper bits.

[0067] The circular buffer outputs of the TX FIR filter are set to zeroafter a reset. This allows null information to be processed until new TXsymbol values are loaded. I data is first processed followed by Q data.

[0068] The TX Clock interrupt signal only occurs during a TX slot. Theprocessor does not know when a TX slot begins or ends except byresponding to this interrupt. The signal has an active low duration ofone 3.2 MHz clock cycle to guarantee that the interrupt is not activeonce it has been serviced. The TX Clock interrupt occurs every othersymbol time (16 KHz/2).

[0069] The RX Clock interrupt occurs for a full frame. The processorchip 12 masks out this interrupt by using the RX Slot marker as a mask.The RX Clock interrupt has an active low duration of one 3.2 MHz clockcycle.

[0070] The RX Start of Slot interrupt occurs every 11.25 milliseconds,and has an active low duration of one 3.2 MHz clock cycle.

[0071] Each interrupt signal is forced to an inactive high state uponreset.

[0072] The codec timing module 44 generates timing strobes and sends thenecessary clock signal via lines 29 to the SLIC and codec circuit 11 tocause 8 bits of data to be transferred between the codec and processorat an 8 KHz rate. The codec 11 receives and transmits 8 bits of dataevery 8 KHz. The codec timing module 44 sends a codec clock signal online 29 a and a codec sync signal on line 29 b. The codec clock signalon line 29 a is generated at a rate of 1.6 MHz by dividing the advanced3.2 Mhz clock by two. An 8 KHz pulse of one 3.2 MHz period is receivedfrom the RX timing circuit 39 and is reclocked to occur for one 1.6 MHzperiod, and thus is guaranteed to occur with respect to the 1.6 MHzclock rising edges. With these two signals, transfer of PCM data betweenthe codec 11 and the processor chip 12 is accomplished. This allows thesubscriber PCM data to be synchronized to the base station PCM data.

[0073] The ringer control module 45 responds to a ring enable controlsignal originating in the processor chip 12 and provided from thecontrol and status register 36 on internal bus 48 by generating a 20 Hzsquare wave signal on line 31 a and two 80 KHz phase control signals,PHASEA on line 31 b and PHASEB on line 31 c and sending these signals tothe ringer circuit 21. The 20 Hz square wave signal on line 31 acontrols the polarity of the ringer voltage provided by the ringercircuit 21 to the telephone interface circuit 10. The 80 KHz phasesignals on lines 31 b and 31 c control the pulse width modulated powersource in the ringer circuit 21.

[0074] A reset or a SLIC ring command signal on line 29 c from the SLICportion of the SLIC and codec circuit 11 turns off or overrides thesesignals on lines 31 a, 31 b, and 31 c after the ring enable signaloriginating in the processor chip 12 has turned them on. This ensuresthat the ringer is off if a reset occurs or the telephone hand set istaken off hook.

[0075] Since the ringer circuit 21 generates a high voltage anddissipates much power, this voltage is not generated except whenrequested by the processor chip 12.

[0076] The external address decoding module 37 generates chip selectsonto the processor bus 25 that are used by the processor chip 12 toaccess the DIF chip 17, the UART hardware, and the slow memory EPROMs 14in separate distinct address segments. The processor chip 12 provideseight MSB address lines, data space, and program space signals. Theseare decoded to generate the appropriate chip selects.

[0077] The watchdog timer module 38 generates a 50 millisecond hardwarereset pulse on line 51, which resets all FIR chip 16 modules and allsubscriber unit modules in FIG. 1. The watchdog timer module 38generates a pulse if it is not reset within a 512 millisecond period bythe Watchdog strobe signal provided on bus 48 by the control and statusregisters 36.

[0078] The DIF chip 17 is interfaced to the processor chip 12 by theprocessor bus 25, to the FIR chip 16 by lines 23 and 24, to the DAC 18by line 71 and to an oscillator in the radio 20 by line 72.

[0079] The oscillator in the radio 20 provides a 21.76 MHz master clocksignal on line 72 to the DIF chip 17.

[0080] Referring to FIG. 3, the DIF chip 17 includes a clock generator60, a processor decoding module 61, a FIR chip interface module 62, aninterpolator 63, a control register 64, tuning registers 65, a DDS phaseaccumulator 66, a DDS SIN and COS generation module 67, a modulator 68and a noise shaper 69. In combination the DDS phase accumulator 66 andthe DDS SIN, COS generator 67 constitute a direct digital synthesizer(DDS) for digitally synthesizing a digital intermediate frequencysignal.

[0081] The DIF chip 17 is an ASIC chip, which is mapped as processordata memory.

[0082] The DIF chip 17 operates in one of two operating modes, amodulated carrier generation mode, and a pure carrier mode. In themodulated carrier generation mode, baseband data is input in the I,Qdomain and this data is used to modulate the pure carrier generated bythe DDS function of the DIF chip 17. In the Pure Carrier Generationmode, the baseband data inputs are ignored and an unmodulated carrierfrom the DDS is provided to the DAC 18.

[0083] The clock generator 60 generates all timing and clocks within theDIF chip 17 and also generates the 3.2 MHz clock signal and the advanced3.2 MHz clock signal that are provided to the FIR chip 16 on lines 23 aand 23 b. The two primary timing signals used within the DIF chip 17 area 21.76 MHz clock and a 2.56 MHz interpolation gate signal. The 3.2 MHzclock is used internally to shift I and Q data on line 24 a from the FIRchip 16 into the FIR interface module 62.

[0084] The clock generator 60 buffers the 21.76 MHz clock received online 72 from the oscillator in the radio 20 and provides a buffered21.76 clock signal on line 71 a. Such buffering is done to providesufficient drive capability for internal functions and to minimize clockskew. The buffered 21.76 MHz clock also provides a clock for the DAC 18and other external circuitry.

[0085] The clock generator 60 provides the 3.2 MHz clock signal bydividing the 21.76 MHz clock by 6 and by 8 in the following sequence:6-8-6-8-6, which thereby results in an average divisor of 6.8(21.76÷6.8=3.2). The effect of this per cycle variation is a minimumperiod of 276 ns and a maximum period of 368 ns. An advanced version ofthe 3.2 MHz clock signal is also generated as the advanced 3.2 MHz clocksignal on line 23 b. Both clocks are identical with the exception thatthe ROM deselect signal on line 23 b leads the 3.2 MHz clock signal online 23 a by one 21.76 MHz clock cycle.

[0086] The clock generator 60 provides the 2.56 MHz gate signal oninternal line 74 by dividing the 21.76 MHz clock by 8 and 9 in an evensequence (8-9-8-9- . . . ), which thereby results in an average divisorof 8.5 (21.76÷8.5=2.56 MHz). This signal is used by the interpolator 63and the modulator 68.

[0087] The processor decoding module 61 allows the processor to controlall internal functions of the DIF chip 17. The processor decoding module61 decodes processor addresses and processor strobes received from dataspace on the processor bus 25 to provide internal write strobes, whichare provided on internal bus 76 to the control register 64 and thetuning registers 65 to enable the processor chip 12 to write control andconfiguration data. Only one output from the processor decoding module61 is active at any given time. The processor addresses determine whichoutput is generated. If a function within the DIF chip 17 address spaceis chosen, a chip select signal on line 24 c from the FIR chip 16becomes active.

[0088] The FIR interface module 62 receives the I and Q samples from theFIR chip 16 on line 24 a in a serial format and converts them into10-bit parallel format in which they are provided to the interpolatormodule on line 77. The I,Q gate signal on line 24 b from the FIR chip 16is used to distinguish the I data from the Q data. The FIR interfacemodule 62 also subtracts previous I and Q samples from current samplesto form a ΔI and ΔQ samples which are then shifted right 4 places (÷16)to form the correct increment for the interpolator module on line 78.Since the FIR interface module 62 supplies data to the interpolator 63,a sync signal is sent by the FIR interface module 62 to the clockgenerator 60 to synchronize the 2.56 MHz gate pulse provided on line 74.

[0089] The interpolator 63 accumulates the ΔI,Q at a 160 KHz×16=2.56 MHzrate and provides interpolated I and Q samples to the modulator 68 onlines 80 and 81 respectively. The interpolator 63 performs a ×16 linearinterpolation in order to reduce the 160 KHz sampling spurs present inthe baseband data received from the FIR chip 16.

[0090] The interpolator 63 successively accumulates the ΔI and ΔQsamples to generate an output at a 2.56 MHz rate. At the end of anaccumulation cycle (16 iterations), the output of the interpolatorshould be equal to the current I and Q samples. This is critical sincethe next accumulation cycle starts its cycle with the current data. Toensure that the data is correct, during the last accumulation cycle thecurrent I and Q data are input directly to the interpolator outputregister in place of the output of the adder (which should have the samedata).

[0091] The control registers 64 are used to control and configure theDIF chip 17 and to select the operating modes. All of the controlregisters 64 are loaded by the processor chip 12 via the processor bus25.

[0092] There are three control registers 64. The first control registerregisters a CW MODE signal, an AUTO TUNE H-L signal, and an AUTO TUNEL-H signal. The second control register registers a SIGN SELECT signal,an OUTPUT CLOCK PHASE SELECT signal, an INTERPOLATOR ENABLE signal, aSERIAL PORT CLOCK SELECT signal, a SERIAL/PARALLEL MODE SELECT signaland a QUADRATURE ENABLE signal. The control functions associated withthese signals are described later at the conclusion of the descriptionof the other modules of the DIF chip 17.

[0093] The third control register enables and specifies the coefficientsfor the noise shaper 69.

[0094] There are three 8-bit tuning registers 65 for storing 24 bits ofphase increment data to specify the frequency of the DDS. This providesa 24-bit tuning word which allows a frequency resolution of (samplerate)/2₂₄=21.76 MHz/2₂₄=1.297 Hz. The output frequency of the DDS isequal to the resolution multiplied by the 24-bit tuning word.

[0095] The tuning registers 65 are loaded by the processor chip 12 viathe processor bus 25. The tuning word is double buffered by the tuningregisters 65 so that the processor chip 12 can write data to theseregisters freely without affecting the current DDS operation.

[0096] The tuning word in loaded from buffer tuning registers intooutput tuning registers whenever a TUNE command is issued. The TUNEcommand is synchronized to the 21.76 MHz clock to provide a synchronoustransition.

[0097] The DDS phase accumulator 66 performs a modulo 2 ₂₄ accumulationof the phase increment provided on line 82 by the tuning registers 65.The output of the phase accumulator 66 represents a digitized phasevalue which is provided on line 83 to the DDS SIN and COS generator 67.The DDS SIN and COS generator 67 generates a sinusoidal function. A DDSworks on the principle that a digitized waveform may be generated byaccumulating phase chances at a higher rate.

[0098] The tuning word, which will be different for different subscriberunits, represents a phase change to the phase accumulator 66. The outputof the accumulator 66 can range from 0 to 2₂₄−1. This intervalrepresents a 360 degree phase change. Although the accumulator 66 worksin standard binary, this digitized phase representation can be input toa waveform generator to produce any arbitrary waveform. In the DIF chip17, the DDS SIN and COS generators 67 produce SIN and COS functions onlines 84 and 85 respectively.

[0099] The period of the waveform function is based on the time requiredto perform a summation to the accumulator upper limit (2₂₄−1). Thismeans that if a large phase increment is provided, then this limit willbe reached sooner. Conversely, if a small increment is given then alonger time is required. The phase accumulator 66 performs a simplesummation of the input phase increment and can be represented by thefollowing equation: $\begin{matrix}{\varphi_{T} = {\sum\limits_{i = 1}^{n}\varphi_{inc}}} & {{Equation}\quad 1}\end{matrix}$

[0100] Where n is the number of iterations, and Φ_(inc) is simply thedata provided on line 82 from the tuning registers 65.

[0101] In the embodiment of the DIF chip 17 described herein, the valueof .phi..sub.T is constrained by the accumulator length to be a maximumof 2.sup.24. Therefore the current phase may be described as:

Φ_(t)=(Φ_(t−1)+Φ_(inc))modulo 2²⁴  Equation 2

[0102] Since the accumulation clock is fixed to be the master 21.76 MHzinput clock this results in a complete cycle taking2.sup.24/.phi..sub.inc iterations at a per iteration period of 1/21.76MHz. So the entire cycle takes the following amount of time:$\frac{2^{24}}{21.76\quad {MH}_{Z}\quad \bullet \quad \varphi_{inc}}$

[0103] Since this period represents a 360 degree cycle, the reciprocalof this expression represents a frequency. The DDS frequency istherefore $\begin{matrix}{f_{DDS} = \frac{21.76\quad {MH}_{Z}\quad \bullet \quad \varphi_{INC}}{2^{24}}} & {{Equation}\quad 3}\end{matrix}$

[0104] In the DDS SIN, COS generation module 67, the SIN and COSwaveforms are generated so a complex mixing may be performed in themodulator. Each is generated by two lookup tables representing a coarseand fine estimate of the waveform. The two values are added to formcomposite 12-bit signed 2's complement SIN and COS data output signalson lines 84 and 85. The lookup tables are implemented in ROM's that areaddressed by the fourteen most significant bits of the signal on line 83from the DDS phase accumulator 66.

[0105] It is desired to have as much phase and amplitude resolution asis practical. In the DIF chip 17 design, 14 bits of phase input and 12bits of amplitude data output are provided in the waveform generationsection. If a “brute-force” approach were taken to generate this datathen very large tables would be needed to generate all possible phaseand amplitude values (e.g. 16K words×12 bits each). To minimize thetable size, the DIF chip 17 makes use of quadrant symmetry andtrigonometric decomposition of the output data.

[0106] Since SIN and COS waveforms have quadrant symmetry, the two mostsignificant bits of the phase data are used to mirror the singlequadrant data around the X and Y axis. For the SIN function theamplitude of the wave in the π to 2 π interval is just the negative ofthe amplitude in the 0 to or π interval. For the COS function theamplitude of the wave in the π/2 to 3 π/2 interval is just the negativeof the amplitude in the 3 π/2 to π/2 interval. The two MSBs of the phaseaccumulator specify the quadrant (00→1, 01→2, 10→3, 11→4). For the SINfunction, the MSB of the phase data is used to negate the positive datagenerated for the first two quadrants. For the COS function, an XOR ofthe two phase data MSBs is used to negate the positive data generatedfor quadrants 1 and 4.

[0107] The above technique reduces memory requirements by a factor of 4.This still results in a memory requirement of 4K words×12 bits To reducethe table sizes further, a trigonometric decomposition is performed onthe angles. The following trigonometric identity is used:

sin Θ.=sin (Φ₁+Φ)=sin Φ₁ cos Φ₂+sin Φ₁ cos Φ₁  Equation 4

[0108] Letting Φ₂<<. _(Φ1) leads to the complete approximation asfollows:

sin θ=sin Φ₁+sin Φ₂ cos Φ₁  Equation 5

[0109] It is not necessary to use all bits of Φ₁ when computing thesecond term of the equation so Φ₁ is a subset of Φ₁. To generate the COSfunction, the same approximation may be used since

cos θ=sin (θ+π/2)  Equation 6

[0110] This results in a modification of the Φ₁ & {circumflex over (Φ)}₁variables when computing the COS function. The data stored in the COSROMs will incorporate this angle modification so no changes to the phasedata are required.

[0111] The modulator 68 mixes the interpolated I and Q samples on lines80 and 81 with the digital intermediate frequency signal represented bythe complex SIN and COS function data on lines 84 and 85 to produce amodulated digital intermediate frequency signal on line 87.

[0112] The interpolated I,Q samples and DDS output are digitally mixedby two 10×12 multipliers. The outputs of the mixing process are thensummed by a 12 bit adder to form a modulated carrier. It is possible toalter the operation of the modulator 68 by forcing the I input to allzeroes and the Q input to all ones. The effect of this is that onemultiplier will output all zeroes and the other will output the signalfrom the DDS SIN, COS generator 67 only. The sum of these two signalsyields an unmodulated digital intermediate frequency signal.

[0113] The modulator 68 creates a modulated digital intermediatefrequency signal on line 87 according to the following equation:

f(t)=I·COS(Φ(t))+Q·SIN(Φ(t))  Equation 7

[0114] The 12-bit output of the DDS SIN and COS generator 67 ismultiplied by the 10 bit interpolated I and Q samples from theinterpolator 63 to generate two 12-bit products. The two products arethen added (combined) to generated a 12-bit modulated output on line 87.

[0115] Since both the I multiplier and the Q multiplier generate 12-bitproducts, it is possible that an overflow could occur when their outputsare combined. Therefore it is necessary to ensure that the magnitude ofthe vector generated by I and Q never exceeds 1 (assuming |I|, |Q| arefractional numbers ≦1). If this is not ensured then an overflow of themodulator adder is possible.

[0116] The noise shaper 69 provides a filtered modulated or unmodulateddigital intermediate frequency signal on line 71 to the DAC 18. Thenoise shaper 69 is designed to decrease the amount of noise power in theoutput spectrum caused by amplitude quantization error.

[0117] The noise filter 69 works on the fact that the quantization noiseis a normal random process, and the power spectral density of theprocess is flat across the frequency band. The desired output signal isoverlayed on top of this quantization noise floor. The noise shapingdevice is a simple multitap finite impulse response (FIR) filter. Thefilter creates a null which decreases the quantization noise power in acertain part of the frequency band. When the desired signal is overlayedon the filtered noise spectrum, the effective SQNR increases.

[0118] The FIR filter transfer function is given by

H(z)=1+bz ⁻¹ −z ⁻²  Equation 8

[0119] A two adder stage creates a second tap value of b in the range of+1.75 to −1.75 (in binary weights of 0, 0.25, 0.50, 1.0) that will movethe zero of the filter across the output frequency band, so that it maybe placed as near as possible to the desired output frequency formaximum SQNR performance.

[0120] The null frequency can be computed by solving for the roots ofthe above equation in the z-plane. The roots are a complex conjugatepair that reside on the unit circle. The null frequency is given by therelation: $\begin{matrix}{f_{null} = {\frac{\Theta}{360{^\circ}}\quad \bullet \quad f_{sampling}}} & {{Equation}\quad 9}\end{matrix}$

[0121] where Θ is the angle of the root in the upper half plane. Theconjugate root will provide a null reflected around the Nyquistfrequency.

[0122] Table 2 lists null frequencies generated by the binary weightedsecond tap. Let b3, b2, and b1 correspond to the weights 1.0 0.5 0.25, a“+” symbol means the tap is equal to its weight, a “−” symbol means thatthe tap is equal to the negative of its weight, and ‘0’ means that thetap has no weight. Some of the null frequencies are equal to those ofother combinations, simply because the possible combinations sometimesoverlap (e.g. 1.0+0.5-0.25=1.0+0.0+0.25). f_(sample) is 1.00. TABLE 2 b3b2 b1 f(null) f(alias) 0 0 0 0.250 0.750 0 0 − 0.269 0.731 0 0 + 0.2300.770 0 + 0 0.210 0.790 0 + + 0.188 0.812 0 + − 0.230 0.770 0 − 0 0.2900.710 0 − + 0.269 0.731 0 − − 0.312 0.688 + 0 0 0.167 0.833 + 0 − 0.1880.812 + 0 + 0.143 0.857 + + 0 0.115 0.885 + + + 0.080 0.420 + + − 0.1430.857 + − 0 0.210 0.790 + − + 0.188 0.812 + − − 0.230 0.770 − 0 0 0.3330.667 − 0 − 0.357 0.643 − 0 + 0.312 0.688 − + 0 0.290 0.710 − + + 0.2690.731 − + − 0.312 0.688 − − 0 0.385 0.615 − − + 0.357 0.643 − − − 0.4200.580

[0123] All timing is derived from the 21.76 MHz clock signal on line 71a.

[0124] The functions associated with the signals in the controlregisters 64 are now described.

[0125] When the CW MODE signal is set, the I input to the respectivemultiplier in the modulator 68 is forced to all zeroes, and thecorresponding Q input forced is to all ones. The net effect is that anunmodulated carrier will be generated. This function is double bufferedand the loaded data will not become active until a TUNE command isissued.

[0126] The INTERPOLATOR ENABLE signal enables the ×16 interpolator onthe I,Q samples. If the INTERPOLATOR ENABLE signal is not set then theI,Q data is input directly to the multiplier.

[0127] External memory required for the operation of the processor chip12 is provided by a fast memory 13 and a slow memory 14. The fast memory13 is accessed by an address decoder 15. The fast memory 13 is a cachememory implemented in a RAM having zero wait states. The slow memory 14is a bulk memory that is implemented in an EPROM, having two waitstates. The slow memory 14 is coupled to the processor chip 12 forstoring processing codes used by the processor chip 12 when said codesneed not be operated with zero wait states; and the fast memory iscoupled to the processor chip 12 for temporarily storing processingcodes used by the processor chip 12 when said codes are operated withzero wait states. When procedures must be run with zero wait states, thecode can be uploaded from the slow memory 14 to the fast memory 15 andrun from there. Such procedures include the interrupt service routines,symbol demodulation, RCC acquisition, BPSK demodulation, and voice anddata processing.

[0128] The processor chip 12 includes a single model TMS320C25 digitalsignal processor, which performs four main tasks, a subscriber controltask (SCT) 91, channel control task (CCT) 92, a signal processing task(SPT) 93, and a modem processing task (MPT) 94, as shown in FIG. 4.These four tasks are controlled by a supervisor module 95. The SCT dealswith the telephone interface and the high-level call processing. The CCTcontrols the modem and RELP operation and timing, and performspower-level and TX timing adjustments according to requests from thebase station. The SPT performs the RELP, echo cancellation and tonegeneration functions. The supervisor calls these four tasks sequentiallyand communicates with them via control words.

[0129] The SCT 91 provides the high level control function within thesubscriber unit and has three fundamental modes of operation: idle,voice and abort.

[0130] The SCT enters Idle Mode after power up and remains in that stateuntil an actual voice connection is made. While in the Idle Mode, theSCT monitors the subscriber telephone interface for activity andresponds to base station requests received over the radio Controlchannel (RCC).

[0131] The primary function of the SCT is to lead the Subscriber Unitthrough the setup and teardown of voice connections on a radio channel.Before the unit can set up any kind of call, however, it must find thecorrect base station. The SCT determines which RCC frequency to use, andsends the frequency information to the CCT. A description of theinitialization of a communication channel between the subscriber unitand the base station is contained in U.S. patent application Ser. No.07/070,970 filed Jul. 8, 1987 now U.S. Pat. No. 4,811,420.

[0132] Once the subscriber unit has gained RCC sync, it can set up acall by exchanging messages over the RCC with the base station, and bymonitoring and setting hardware signals on the telephone interface. Thefollowing walk through briefly describe the events that take placeduring call setup.

[0133] Normal call setup for call origination begins with the subscribertaking the handset off hook to initiate a service request. The SCT sendsa CALL REQUEST message to the base station. The SCT receives a CALLCONNECT message. The SCT signals the CCT to attempt sync on the voicechannel assigned via the CALL CONNECT message. The CCT attains sync onthe voice channel. The subscriber receives a dial tone from the centraloffice. Call setup is complete. The central office provides theremaining call termination support.

[0134] Normal call setup for call termination takes place as follows:The SCT receives a PAGE message from the base station. The SCT replieswith a CALL ACCEPT. The SCT receives a CALL CONNECT message. The SCTsignals the CCT to attempt sync on the voice channel assigned via theCALL CONNECT message. The CCT attains sync on the voice channel. The SCTstarts the Ring Generator to apply ring to the local loop. Thesubscriber takes the hand set off hook. The ringing is stopped. Thevoice connection is complete.

[0135] The SCT implements the call setup and teardown operations as afinite state machine.

[0136] If a voice channel seizure is successfully completed, the SCTswitches to the voice mode and performs a very limited set of supportfunctions. SCT processor loading is kept to a minimum at this time togive the RELP speech compression, echo cancellation and modem processingalgorithms maximum processor availability.

[0137] The SCT enters the abort mode as a result of an unsuccessful callorigination attempt or an unexpected call teardown sequence. During theabort mode, a reorder is sent to the handset. The SCT monitors thesubscriber telephone interface for a disconnect (extended on-hook), atwhich time the subscriber unit enters the idle Mode. Base stationrequests received over the radio control channel (RCC) are rejecteduntil the disconnect is detected.

[0138] The CCT 92 operates as a link level channel controller in thebaseband software. The CCT has three fundamental states: RCC operation,refinement, and voice operation.

[0139] At power up, the CCT enters the RCC operation state to search forand then support the RCC channel. The RCC operation includes thefollowing functions: AM hole control; monitoring sync and modem taskstatus; radio channel timing adjustment; RX RCC message filtering; TXRCC message formatting; monitoring the PCM buffer I/O; and linkinformation processing.

[0140] After a voice connection is established, the CCT enters therefinement state to fine tune the modem's fractional timing. Refinementincludes the following functions; interpreting and responding torefinement bursts; creating and formatting TX refinement bursts;forwarding messages to the SCT as appropriate; monitoring the modemstatus; and monitoring the PCM buffer I/O.

[0141] Following Refinement, the CCT begins voice operation, whichincludes the following functions: code word signalling support; dropoutrecovery; monitoring sync and modem status; and monitoring the PCMbuffer I/O.

[0142] The CCT 92 has three fundamental states of operation: idle,refinement and voice. The following is a walk through of the statetransitions involved in CCT operation.

[0143] After a reset the CCT enters the idle state and remains inactiveuntil given channel assignment instructions by the SCT. The SCT providesthe CCT with a frequency upon which to search for the radio controlchannel (RCC). The CCT then instructs the MPT to synchronize thereceiver to the given frequency and to search for an AM hole. Failure todetect an AM hole within a predetermined time period causes the CCT torequest another frequency upon which to search from the SCT. Thiscontinues indefinitely until the AM hole detection is successful.

[0144] Following a successful AM hole detection, the CCT begins to checkreceived data for the unique word. A small window around the nominalunique word position is scanned since the AM hole detection process maybe off by a few symbol times. Once the unique word is located and theCRC error detection word is verified correct, the exact receive symboltiming can be determined. The TDM framing markers are then adjusted tothe correct alignment and normal RCC support begins. If the unique wordcannot be located, the AM hole detection is considered false and the CCTrequests a new frequency assignment from the SCT.

[0145] During RCC operation the CCT filters received RCC messages. Themajority of the base station's RCC messages are null patterns and theseare discarded after link information is read from the link byte. RCCmessages that contain real information are forwarded to the SCT forprocessing. If RCC synchronization is lost, the CCT again requests a newfrequency from the SCT. The SCT will respond with the correct frequencyaccording to the RCC frequency search algorithm.

[0146] When the SCT initiates a voice call, the CCT is assigned a voicechannel and time slot. The CCT brings the subscriber unit activeaccording to this assignment and begins the refinement process. Duringrefinement, the base and subscriber units transmit a BPSK signalspecifically designed to assist the modem in fractional bit timeacquisition. The base station CCU relays the bit timing offset back tothe subscriber unit as a two's complement adjustment value. The CCTmaintains a time average of these fedback offsets. Once the CCTdetermines that the fractional timing value is within a requiredtolerance, it adjusts the subscriber unit's transmit timing accordingly.The length of the time average is determined dynamically, depending uponthe variance of the fractional time samples. After a timing adjustment,the time average is reset and the procedure is repeated.

[0147] Once the base station detects that the subscriber unit is withinan acceptable timing tolerance, it terminates the refinement process andvoice operation begins. The length of the refinement process isdetermined dynamically, depending upon the success of the subscriberunit's timing adjustments. Power and integer symbol timing are alsomonitored and adjusted as necessary during the refinement process. Ifthe subscriber fails to find the base station's refinement bursts aftera period of time, or if the refinement process cannot yield acceptabletiming, the connection is broken and the CCT returns to RCC operation.

[0148] Following successful refinement, the CCT enters voice operationat the assigned modulation level. The voice operation tasks includecontrolling RELP and MPT operations, establishing voice synchronizationand continuously monitoring the voice code words sent from the basestation. Local loop control changes, signalled via the code words, arereported to the SCT as they occur. Power and fractional timingincremental changes are also determined from the code words. Transmittedvoice code words are formulated by the CCT based upon the local loopcontrol provided by the SCT and the channel link quality reported by themodem. The CCT returns to the RCC when the SCT executes a call teardownsequence.

[0149] If voice synchronization is lost, the CCT initiates a faderecovery operation. After ten seconds of failure to reestablish a goodvoice connection, the CCT informs the SCT of the condition, initiating acall teardown. This returns the CCT to the Idle state.

[0150] During a channel test operation, a voice burst is replaced withchannel test data. When a burst has just been received, it is analyzedfor bit errors. The bit error count is passed to the base station viathe reverse channel bursts.

[0151] The SPT 93 performs all of the digital signal processing (DSP)tasks within the subscriber unit. The various DSP functions are invokedas required, under the control of the supervisor module 95.

[0152] The SPT includes a RELP module. which is executed from a highspeed RAM. The RELP module performs RELP Speech compression andexpansion with echo cancellation. The RELP module transforms 180 byteblocks of 64 Kbps PCM voice data to and from 42 bytes of compressedvoice data using the RELP algorithm.

[0153] The SPT also includes a signal processing control (SPC) module,which determines if tone generation or RELP should be invoked. If RELP,SPC determines whether to call the synthesis or analysis routines. Thesynthesis routine returns a parity error count, which is handled by theSPTCTL routine. If tone generation is required, it determines whether tooutput silence or reorder.

[0154] The SPT is controlled via commands from the SCT and the CCT.These commands invoke and control the operation of the various functionswithin the SPT as they are required by the subscriber unit. RELP andecho cancellation software, for example, are only executed when thesubscriber unit is active on a voice call. Call progress tones aregenerated anytime the subscriber unit's receiver is off hook and RELP isnot active. The tones include silence and reorder. Except for the IDLEmode, the interrupt service routine handling the PCM codec operatescontinuously as a foreground process, filling the circular PCM buffer.

[0155] The control and modem functions are performed in between theanalysis and synthesis processing.

[0156] The MPT 94 demodulation procedure is divided into two procedures:DEMODA & DEMODB, thus allowing the RELP synthesis to be executed on theRX data in buffer A right after the DEMODA procedure is completed. AfterDEMODA all internal RAM variables should be stored in external RAM, thenreloaded to internal RAM before performing DEMODB. This is because RELPuses the internal RAM.

[0157] When the RXCLK interrupt on line 26 c is received by theprocessor chip 12, the MPT causes four received RX data samples to beread and then placed in a circular buffer, for processing by thedemodulation procedure. This allows other tasks to be performed whilereceiving RX samples.

[0158] The MPT receives the RXCLK interrupt signal on line 26 c from theFIR chip 16 every 62.5 .mu.sec during the receive slot. The RXCLKinterrupt signal is masked by the processor chip firmware during idle ortransmit slots.

[0159] The MPT receives the TXCLK interrupt signal on line 26 c from theFIR chip 16 only during the transmit slot. The TXCLK interrupt signaltells the processor chip 12 when to send a new TX symbol to the FIRchip.

[0160] The MPT reads four samples from the RX sample buffer 35 in theFIR chip 16 during each RXCLK interrupt on line 26 c. The MPT resets theinput and output address counters to the buffer at the start of thereceive slot.

[0161] The MPT sends TX symbols to the TX symbol buffer 36 in the FIRchip 16.

[0162] The MPT provides the data to the fractional timing circuit in theRX timing module 39 in the FIR chip 16 that is used to align the RXCLKinterrupt signal on line 26 c with the base station transmission.

[0163] The MPT also synchronizes the DDS frequency to the base stationtransmit frequency.

[0164] Referring to FIG. 5 the MPT includes the following modules: asupervisor module 101, a training module 102, a frequency acquisitionmodule 103, a bit synchronization module 104, a voice demodulationmodule 105, a symbol receive module 106, and a transmit module 107.

[0165] The supervisor module 101 is the MPT task supervisor. It readsthe MPT control word (CTRL0) from the RAM, and calls other routinesaccording to the control word.

[0166] The training module 102 computes a vector of 28 complex FIRfilter coefficients. It is activated in the idle mode after power up andabout every three hours. A training transmitter implemented by the MPTis activated in a loopback mode to send a certain sequence of symbols.This sequence is looped back to a training receiver implemented by theMPT, in a normal mode, in advanced and delayed timing modes, and inupper and lower adjacent channels.

[0167] The training receiver uses the samples of the input waveform tocreate a positive definite symmetric matrix A of order 28. Also a28-word vector V is created from the input samples. The coefficientsvector C is given by:

C=A⁻¹ V  {Eq. 10}

[0168] The B coefficient is then calculated according to the algorithm:B=A⁻¹ given A.

[0169] The training transmitter is activated in the loopback mode totransmit five similar pairs of sequences. Each pair consists of thefollowing two sequences:

[0170] I sequence: 9 null symbols, “i”, 22 null symbols

[0171] Q sequence: 9 null symbols, “j”, 22 null symbols

[0172] The “i” can be any symbol. The “j” is a symbol that differs from“i” by 90 degrees.

[0173] The receiver processing tasks are:

[0174] Adjust the AGC so that the signal peak in the normal mode is 50to 70% of the maximum. The AGC is increased by 23 db for the 4th and 5thmodes.

[0175] Read and store the input samples. The first 32 samples arediscarded and the next 64 samples are stored, for each sequence.

[0176] Build the matrix A(28,28). The following process is done in thenormal mode:

A(I,J)=A(I,J)+ΣX(4N−I)·X(4N−J)  Equation 11

[0177] The addition is for all N that satisfy:

0<=4N−I<64 & 0<=4N−J<64  Equation 12

[0178] For the advanced and delayed sequences, the same process isperformed except that the term resulting from N=8 is not added. In theupper and lower adjacent channel channel sequences, the followingprocess is performed:

A(I,J)=A(I,J)+ΣX(2N−I)·X(2N−J)  Equation 13

[0179] The addition is for all N that satisfy:

0<=2N−I<64 & 0<=2N−J<64  Equation 14

[0180] Create the vector V(1:28) from the samples of the first pair ofsequences:

[0181] Re{V(I)}=X(32−I); where X are samples of the first (I) sequence.

[0182] Im{V(I)}=X(32−I); where X are samples of the second (Q) sequence.

[0183] Find the coefficients vector C by solving the equation:

A×C−V=0  Equation 15

[0184] These processing steps are more fully described in U.S. Pat. No.4,644,561 issued Feb. 17, 1987 to Eric Paneth, David N. Critchlow andMoshe Yehushua.

[0185] The frequency acquisition module 103 is run when receiving thecontrol channel, in order to synchronize the subscriber unit RXfrequency to the base station transmit frequency. This is done byadjusting the DDS CW output until the energies of the received signal'stwo sidebands are equal. Afterwards, the DDS TX frequencies are adjustedaccording to the computed frequency deviation.

[0186] If the procedure fails to achieve frequency sync, an appropriateerror code is placed in the status word.

[0187] The bit synchronization module 104 is run when receiving the RCCand after completing the frequency acquisition. A certain pattern istransmitted in the first 44 symbols in the RCC transmission from thebase station, and this is used by this module to compute the RXCLKdeviation from the correct sampling time. This deviation is used toadjust the RXCLK timing.

[0188] The voice demodulation module 105 is activated to demodulate avoice slot. It is resident in the slow EPROM and its functions aredivided between two procedures DEMODA and DEMODB.

[0189] The DEMODA functions include initializing parameters for thesymbol receive module 106; calling the symbol receive module to processthe received symbols for buffer A; and storing the variables in externalRAM before exiting.

[0190] The DEMOOB functions include loading the variables from externalRAM to internal RAM; calling the symbol receive module to process thereceived symbols for buffer B; and determining link quality and otherinformation after receiving all the symbols in the slot.

[0191] The symbol receive module 106 is uploaded to the RAM when the CCTgoes to the voice mode. It is called by DEMODA or DEMODB to perform thefollowing: (1) read I and Q samples from the circular buffer; (2) FIRfiltering of the I&Q samples; (3) determine the transmitted symbols andand put them in a buffer; (4) execute a phase-lock-loop to synchronizethe DDS to the incoming signal; (5) execute the bit tracking algorithm;(6) AGC calculation; and (7) accumulate data for link qualitycalculation.

[0192] The transmit module 107 includes the interrupt service routinefor the TXCLK interrupt signal received on line 26 c from the FIR chip16, which occurs once per two symbols during a transmit slot. Thefunctions of the transmit module 107 include: (1) unpacking the transmitsymbol from the RELP buffer; (2) performing an inverse GRAY coding onit; (3) adding it to the previous transmitted phase (because of the DPSKtransmission); and (4) sending it to the TX buffer in the FIR chip 16.

[0193] The interface of the MPT to the baseband tasks is accomplishedvia control and status words and data buffers in the shared memory.Procedures requiring fast execution are uploaded into the cache memorywhen needed. These include the interrupt service routines. symboldemodulation, RCC acquisition; and BPSK demodulation.

[0194] The MPT supervisor will not wait for RXSOS to read and decode thecontrol word, but will do that immediately when it is called.

[0195] The TMS320C25 goes to a powerdown mode when executing the IDLEinstruction. In order to conserve power the firmware will be in the idlemode most of the time, when there is no phone call in progress. So aftera reset the supervisor will acquire RCC sync then go to idle mode untila predetermined interrupt causes a corresponding service routine to beexecuted. When operated in the powerdown mode, the TMS320C25 enters adormant state and requires only a fracion of the power normally neededto supply the device. While in powerdown mode, all of the internalcontents of the processor are maintained to allow operation to continueunaltered when the powerdown mode is terminated. Upon receipt of aninterrupt the processor chip 12 terminates the powerdown mode temporallyand resumes normal operation for a minimum time of one main loop cycle.The requirements of the powerdown mode are checked at end of main loopevery time to determine whether or not the subscriber unit to return tothe powerdown mode.

[0196] The slot clock is based on the hardware generated slot timing.When a slot marker triggers an interrupt, the routine increments theclock by one tick. Each clock tick represents 11.25 ms in time.

[0197] The receive and transmit functions of the UART are not interruptdriven, but are controlled by the background software (this controlsprocessor loading and prevents runaway interrupt conditions). The.processing code supports the XON/XOFF protocol by intercepting thesecharacters directly and immediately enabling or disabling UARTtransmission as appropriate. The rate of the receive and transmitoperation is designed to be selective by an external DIP switch device.The typical data reception rate is at 9600 baud. A circular buffer isused to control the UART's transmission. The background softwareperiodically checks the queue and initiates transmission if it is notempty. It does this by sending bytes to the UART one byte at a timeuntil the queue is empty.

[0198] The switch hook is sampled with the TMS320C25 internal timerinterrupt routine. To simulate DC signalling, a 1.5 ms sample period isused. This interrupt is aligned to frame timing at the beginning of eachframe therefore its frequency is phase locked to the base station toprevent underrun or overflow of the switch hook buffer. For eachinterrupt, a bit representing the switch hook detect signal (from theSLIC) is entered in the 60-bit Switch Hook Sample buffer (SSB). The SSBis examined by the SCT once every 45 ms during normal operation. Thisinterrupt is enabled by the software at all times.

What is claimed is:
 1. A system for processing communication signalswhich include a first information signal and a first communicationsignal which carries information contained in digital input symbols, anda second communication signal provides a second information signal, thecommunication signals provided over a plurality of radio frequencychannels within a selected band of radio frequencies, the systemcomprising: a circuit for generating a base radio frequency signal; acircuit for generating a digital intermediate frequency signal such thatthe combination of the digital intermediate frequency signal with thebase radio frequency signal produces a communication signal having afrequency within one of said radio frequency channels selected forcommunication; a circuit for modulating the digital intermediatefrequency signal with the digital input symbols to produce a modulateddigital intermediate frequency signal, and combining the modulateddigital intermediate frequency signal with the base radio frequencysignal to produce the first communication signal; and a circuit forcombining the modulated digital intermediate frequency signal with thebase radio frequency signal to produce the first communication signal.2. The system of claim 1 further comprising a circuit for accumulatingphase increment data to produce digitized phase values and generatingthe digital intermediate frequency signal based on the digitized phasevalues.
 3. The system of claim 1 further comprising a demodulationcircuit, the demodulation circuit using the digital intermediatefrequency signal to demodulate the second communication signal.
 4. Thesystem of claim 3 further comprising a filter, including a noise shapingcircuit, for filtering the digital intermediate frequency prior todemodulating the second communication signal received.
 5. The system ofclaim 1 further comprising: a memory; and the circuit for generating thedigital intermediate frequency signal based on the digitized phasevalues by using predefined values stored in the memory.
 6. The system ofclaim 1 comprising: a lookup table comprising a memory having a set ofpredefined stored values pertaining to the amplitude of a signal for asingle quadrant; a signal generator receiving an input signal andgenerating sine and cosine waveforms utilizing amplitude values obtainedfrom said lookup tables, wherein the input signal includes phase dataand specifies the quadrant and the algebraic sign of the phase data;said signal generator accessing said lookup table differently dependingupon the quadrant and sign of the phase data such that the lookup tableprovides an amplitude value from said set of values based upon the phasedata; and a circuit combining the sine and cosine waveforms to producethe digital frequency.
 7. The system of claim 6 wherein the lookup tableincludes a first table having stored values pertaining to large angleapproximations for coarse resolution frequency adjustment of the digitalfrequency and a second table for stored values pertaining to small angleapproximations for fine resolution frequency adjustment.
 8. The systemof claim 6 wherein trigonometric decomposition is utilized to furtherreduce table size.
 9. The system of claim 1 further comprising thecircuit for modulating the digital intermediate frequency signaltranscoding the first information signal into digital input symbols andmodulating the digital intermediate frequency signal with the digitalinput symbols to produce the modulated digital intermediate frequencysignal.
 10. The system of claim 1 further comprising the circuit forgenerating the digital intermediate frequency signal using predefinedsine and cosine waveform values stored in the memory to providedigitized phase values.
 11. The system of claim 10 further comprisingthe circuit for generating the digital intermediate frequency usingcoarse and fine resolution frequency approximations stored in thememory.
 12. The system of claim 11 further comprising the memoryproviding two lookup tables.
 13. The system of claim 12 furthercomprising the circuit for generating the digital intermediate frequencysignal reducing the table sizes by utilizing quadrant symmetry of thesine and cosine waveform values.
 14. The system of claim 13 furthercomprising the circuit for generating the digital intermediate frequencysignal utilizing trigonometric decomposition, thereby reducing requiredtable sizes.
 15. The system of claim 1 comprising: a lookup tablecomprising a memory having a set of predefined stored values pertainingto the amplitude of a signal for a single quadrant; a signal generatorreceiving an input signal and generating sine and cosine waveformsutilizing amplitude values obtained from said lookup tables, wherein theinput signal includes phase data and specifies the quadrant and thealgebraic sign of the phase data; the lookup table providing anamplitude value from said set of values based upon the phase data; and acircuit combining the sine and cosine waveforms to produce the digitalfrequency.
 16. The system of claim 15 comprising: the lookup tableincluding a first table having stored values pertaining to large angleapproximations for coarse resolution frequency adjustment of the digitalfrequency and a second table for stored values pertaining to small angleapproximations for fine resolution frequency adjustment; and the circuitfor generating the digital intermediate frequency signal utilizingtrigonometric decomposition, thereby reducing required table sizes. 17.The system for processing communication signals which include a firstinformation signal and a first communication signal which carriesinformation contained in digital input symbols, and a secondcommunication signal provides a second information signal, thecommunication signals provided over a plurality of radio frequencychannels within a selected band of radio frequencies, the systemcomprising: means for generating a base radio frequency signal; meansfor generating a digital intermediate frequency signal such that thecombination of the digital intermediate frequency signal with the baseradio frequency signal produces a communication signal having afrequency within one of said radio frequency channels selected forcommunication; means for modulating the digital intermediate frequencysignal with the digital input symbols to produce a modulated digitalintermediate frequency signal; and circuit means for combining themodulated digital intermediate frequency signal with the base radiofrequency signal to produce the first communication signal.
 18. Thesystem of claim 17 further comprising: demodulation means, responsive tothe digital intermediate frequency signal to demodulate the secondcommunication signal; and means for filtering the digital intermediatefrequency by noise shaping prior to demodulating the secondcommunication signal.
 19. The system of claim 17 further comprising: themeans for generating a digital intermediate frequency signal generatingthe digital intermediate frequency signal based on predefined valuesstored in a memory store; demodulation means, responsive to the digitalintermediate frequency signal to demodulate the second communicationsignal; the means for generating the digital intermediate frequencysignal generating the signal based on the digitized phase values byusing by utilizing quadrant symmetry of sine and cosine waveform valuesstored in the memory store; the means for generating the digitalintermediate frequency using coarse and fine resolution frequencyapproximations stored in the memory store.
 20. The system of claim 17comprising: a lookup table comprising a memory having a set ofpredefined stored values pertaining to the amplitude of a signal for asingle quadrant, the lookup table including a first table having storedvalues pertaining to large angle approximations for coarse resolutionfrequency adjustment of the digital frequency and a second table forstored values pertaining to small angle approximations for fineresolution frequency adjustment; means for generating sine and cosinewaveforms utilizing amplitude values and phase data obtained from saidlookup tables; and means for combining the sine and cosine waveforms toproduce the digital frequency.
 21. The system of claim 20 whereintrigonometric decomposition is utilized, thereby reducing table size.